1. Field of the Invention
The present invention relates to a self-oscillation switching power supply, and more particularly, to a switching power supply for outputting a high voltage.
2. Description of the Related Art
A ringing choke converter is widely used as a self-oscillation switching power supply. FIG. 16 is a circuit diagram of a ringing choke converter according to a conventional technique. In FIG. 16, reference numeral 11 denotes a DC power supply circuit which generates a DC voltage of about 120 V by rectifying and smoothing commercial AC electric power. T denotes a transformer with a primary winding Lp, a secondary winding Ls, and a feedback winding Lf. Q1 denotes a switching transistor connected to the DC power supply via the primary winding Lp of the transformer T. The base of the switching transistor Q1 is connected to a starting resistor R1. The base of the transistor Q1 is also connected to the feedback winding Lf via a current limiting resistor R2, a speed-up capacitor C2, and a diode D2. Furthermore, there is disposed a control transistor Q2 between the base and the emitter of the switching transistor Q1. The feedback winding Lf is connected to a time constant circuit 4 comprising a resistor R5 and a capacitor C3 wherein the voltage across the capacitor C3 is applied to the base of the transistor Q2. The secondary winding Ls of the transformer T is connected to a rectifying and smoothing circuit 2 comprising a rectifying diode D1 and a smoothing capacitor C1. The output side of the rectifying and smoothing circuit 2 is connected to a resistance voltage divider comprising resistors R3 and R4, a variable shunt regulator 12, and a light emitting diode of a photocoupler PC. A phototransistor of the photocoupler PC is disposed in the charging path for charging the capacitor C3.
The power supply shown in FIG. 16 operates as follows. If a DC voltage is applied from the DC power supply circuit 11, a small starting current flows into the base of the switching transistor Q1 via the starting resistor R1. As a result, a current flows through the collector of Q1. This causes a reduction in the collector-emitter voltage and thus a voltage is applied between the terminals of the primary winding Lp of the transformer T. In proportion to this voltage, a voltage is induced across the feedback winding Lf. This induced voltage causes a positive feedback current to be supplied to the base of the switching transistor Q1 via the current limiting resistor R2, the speed-up capacitor C2, and the diode D2. As a result, the transistor Q1 is turned on (into a saturated state). In response to the transition of Q1 into the on-state, a DC voltage is applied between the terminals of the primary winding Lp of the transformer T and a current flows through the primary winding Lp. As a result, the transformer is excited. At the same time, a voltage is induced across the feedback winding Lf whereby the capacitance C3 is charged via the resistor R5, the speed-up capacitor C2, the diode D2, and the phototransistor of the photocoupler PC. When the charging voltage across the capacitor C3 reaches a threshold value (about 0.6 V) of the base-emitter voltage of the control transistor Q2, the base and the emitter of the switching transistor Q1 are short-circuited by Q2 and thus the base current of the switching transistor Q1 is cut off. As a result, Q1 quickly turns off. Herein, the switching transistor Q1 is in the on-state during the period from the time at which the capacitor C3 is started to be charged to the time at which the voltage across the capacitor C3 reaches about 0.6 V. If the switching transistor Q1 turns off, the base of the switching transistor Q1 is reverse-biased to a negative value by a voltage induced across the feedback winding Lf. At the same time, the capacitor C3 is forced to be discharged (reversely charged) by the feedback winding Lf via the resistor R5. As a result, the base of the control transistor Q2 is reverse-biased to a negative voltage. Thus, the transistor Q2 is maintained in the off-state until the excited energy of the transformer T is entirely released from the secondary winding Ls. If the excited energy of the transformer T is entirely released, the voltage induced across the feedback winding Lf disappears quickly. However, a ringing voltage (kick voltage) is generated by the leakage inductance and the distributed capacitance of the transformer T whereby the base of the switching transistor Q1 is forward-biased and thus the switching transistor Q1 turns on again. The above-described turning on and off occurs periodically and the oscillatory operation grows into a continuous oscillation.
If the output voltage appearing between the terminals of the rectifying and smoothing circuit 2 is denoted by Vout, the current flowing through the load is denoted by Iout, the inductance of the primary winding Lp is denoted by Lp, and the peak value of the collector current of the switching transistor Q1 is denoted by Icp, then the output voltage Vout can be approximated by the following equation. EQU Vout=(Lp.multidot.Icp.sup.2)/(2Iout) (1)
Furthermore, if the on-period of the switching transistor Q1 is denoted by ton, and the voltage applied between the terminals of the primary winding Lp during the on-period is denoted by Vin, then Icp is given by the following equation. EQU Icp=(Vin/LP)ton (2)
According to the relationships given by equations (1) and (2), it is possible to detect the output voltage and control the current of the phototransistor of the photocoupler PC thereby controlling the on-period ton of the switching transistor Q1 so that the output voltage Vout is maintained at a fixed value.
In the conventional self-oscillation switching power supply shown in FIG. 16, a step-down transformer is employed as the transformer T, and the output voltage Vout is limited to a rather low voltage such as 5 V. Using the circuit configuration shown in FIG. 16, it is possible to produce a power supply capable of generating a high voltage by increasing the ratio of the number of turns of the secondary windings Ls to that of the primary winding Lp of the transformer T. However, such a power supply will have the following problems.
FIG. 17 is a circuit diagram of a transformer wherein Cs denotes a distributed capacitance appearing across the secondary winding Ls, and Cps denotes a distributed capacitance appearing between the primary winding Lp and the secondary winding Ls. Cpp denotes a capacitance which is placed between the terminals of the primary winding Lp to equivalently replace the distributed capacitances Cs and Cps. For example, in copying machines and page printers of the electrophotographic type, a power supply is required to convert a DC input voltage of a few ten volts to a DC or AC voltage of an increased value of about a few hundred or thousand volts. To meet such a requirement, the high-voltage transformer should have an extremely high ratio of the number of turns of the secondary winding Ls to that of the primary winding Lp. If the number of turns of the primary winding Lp is denoted by Np, the number of turns of the secondary winding Ls is denoted by Ns, and the values of the distributed capacitances Cs and Cps are denoted by Cs and Cps, then the equivalent distributed capacitance Cpp between the terminals of the primary winding can be approximated by the following equation. EQU Cpp=(Cs+Cps).times.(Ns/Np).sup.2 (3)
This means that the capacitance Cpp of the high-voltage transformer becomes extremely high compared to that of the low-voltage transformer. Furthermore, if the inductance of the primary winding Lp is denoted by Lp, then the intrinsic resonance frequency fo of a parallel combination of the inductance Lp of the primary winding and the equivalent capacitance Cpp across the primary winding is given by the following equation. EQU fo=1/(2.pi.(Lp.multidot.Cpp).sup.1/2) (4)
From this equation, it can be seen that the resonance frequency fo of the high-voltage transformer is lower than that of the low-voltage transformer.
If the transformer T shown in FIG. 16 is replaced with the high-voltage transformer shown in FIG. 17, the operation becomes very different in that the high-voltage transformer has a free oscillation at a resonance frequency fo determined by equation (4) during the period from the instant at which the switching transistor Q1 turns off to the instant at which it turns on again. In the conventional low-voltage switching power supply shown in FIG. 16, the oscillation frequency varies to a great extent depending on the output power consumption. More specifically, as the output power consumption decreases, it becomes possible to excite the low-voltage transformer T in a shorter on-period. As a result, the oscillation frequency tends to become higher. The capacitance Cpp in equation (3) for the low-voltage transformer T is very small, and the intrinsic resonance frequency of the transformer is very high, and thus oscillation even at a few hundred kHz is possible. In contrast, the high-voltage transformer has a very low intrinsic resonance frequency fo, as described above, and therefore, it is difficult to achieve oscillation at a frequency higher than the intrinsic resonance frequency even when there is no load which consumes the output power.
FIGS. 18A to 18C illustrate the waveforms of the collector-emitter voltage of the switching transistor for various output voltages (currents). In order to vary the output voltage (current) over a wide range using the high-voltage transformer, it is required to vary the amplitude of the collector-emitter voltage Vce of the switching transistor Q1 while maintaining the repetition frequency at the resonance frequency fo as shown in FIG. 18. To this end, the switching transistor Q1 should operate over a wide operating range including both in a saturation region and in an unsaturated region. When the switching transistor Q1 operates in the unsaturated region, the positive feedback voltage appearing across the feedback winding Lf has a sinusoidal voltage waveform having a decreasing amplitude, as shown in FIG. 18C. In this operation region, the on-period of the switching transistor Q1 becomes short and the peak value of the positive feedback voltage across the feedback winding Lf decreases. However, it is still required to reduce the on-period of the switching transistor Q1 in response to the feedback signal given via the photocoupler PC and thus the capacitor C3 is charged via the phototransistor, which is in the initial saturated state, of the photocoupler PC. During this process, the feedback winding Lf is substantially short-circuited by the path including the diode D2, the phototransistor of the photocoupler PC, and the capacitor C3. As a result, the capacitor C3 is further charged without supplying a positive feedback current from the feedback winding into the base of the switching transistor Q1. As a result, a delay occurs in the supply of the base current from the positive feedback winding Lf to the switching transistor Q1. This causes the control transistor Q2 to turn on before the switching transistor Q1 turns on, and thus the switching transistor Q1 operates in an intermittent manner in which the switching transistor Q1 becomes alternately saturated and unsaturated. Therefore, it is impossible to achieve stable control and the output voltage includes large ripples.
Leakage inductance of the transformer is another problem which occurs when the low-voltage transformer T shown in FIG. 16 is replaced with the high-voltage transformer shown in FIG. 17. FIG. 19 illustrates an equivalent circuit of the high-voltage transformer and the switching transistor. In FIG. 19, L1 and L2 denote leakage inductance and Lp denotes the exciting inductance of the primary winding. Cpp denotes the equivalent primary-side distributed capacitance shown in FIG. 17. If the inductance component of the leakage inductances L1 and L2 is denoted by L.sub.1e, then the series resonance frequency fo' is given by the following equation . EQU fo'=1/(2.pi.(L.sub.1e .multidot.Cpp).sup.1/2) (5)
As described above, the high-voltage transformer has a very large equivalent primary-side capacitance Cpp and thus a rather low series resonance frequency fo' determined by equation (5). Although the series resonance frequency fo' varies depending on the leakage inductance L.sub.1e, a typical value is of the order of 6 to 10 times the parallel resonance frequency given by equation (4). That is, the series resonance frequency is rather close to the parallel resonance frequency. Therefore, if the high-voltage transformer with such characteristics is applied to the circuit shown in FIG. 1, a ringing component is superimposed on the collector-emitter voltage Vce of the switching transistor Q1 as shown in FIGS. 20A to 20C. The high-voltage power supply is required to have the capability of varying the output voltage (current) over a wide range. To meet this requirement, if the voltage induced across the secondary winding Ls of the high-voltage transformer is varied over a wide range by adjusting the on-period of the switching transistor Q1, the collector-emitter voltage of the switching transistor Q1 varies as shown in FIG. 20. As can be seen from FIGS. 20A to 20C, the series resonance frequency component described in equation (5) becomes more dominant with the reduction in the on-period of the switching transistor Q1.
In the conventional circuit shown in FIG. 16, the capacitor C3 is discharged by the voltage induced across the feedback winding Lf and charged by the voltage induced across the feedback winding Lf and the current passing through the phototransistor of the photocoupler PC. In such a circuit configuration, when the circuit is in the oscillating state shown in FIG. 20C, the time constant circuit 4 responds to the series resonance frequency fo' because the time constant circuit 4 is formed with passive elements. As a result, the control transistor Q2 also responds to the series resonance frequency component. As a result, the switching transistor Q1 operates not in the parallel resonance mode which is the right operation mode in which the transistor Q1 should be operated, but in the series resonance mode. This causes an unstable circuit operation such as intermittent oscillation. Furthermore, because the switching transistor Q1 turns on and off at a high frequency, the switching loss increases and thus it becomes necessary to employ a larger-size heat sink.
Furthermore, since the high-voltage transformer has, as described above, an extremely high equivalent primary-side distributed capacitance Cpp compared to the low-voltage transformer, a large excess current flows when the switching transistor turns on. The waveform of the collector current of the switching transistor Q1 and some other related waveforms are shown in FIGS. 21A to 21D. FIGS. 22A and 22B are waveforms illustrating the relationship among the collector voltage and current and the base voltage and current of the switching transistor Q1. In FIGS. 21A to 21D, V.sub.Lp is the waveform of the voltage applied to the primary winding, Ic' is the current flowing through the equivalent primary-side distributed capacitance Cpp, I.sub.Lp is the primary winding current, and Ic is the collector current of the switching transistor Q1. When the switching transistor Q1 turns on, an excessively large value of current Ic' flows into Cpp thereby initially charging Cpp. After completion of the charging, oscillation occurs due to the resonance between Cpp and the leakage inductances (L1, L2). The amplitude of the oscillation decreases with time. The sum of Ic' and I.sub.Lp flows through the collector of the switching transistor Q1, and thus an initial current of Ic with an excessively large value flows as shown in FIG. 21D. After that, the waveform of Ic includes a component increasing at a rate V.sub.Lp /L.sub.p and a ringing component. This ringing component superimposed on the collector current Ic adversely affects the control operation of the circuit based on the technique in which the output is stabilized by controlling the on-period of the switching transistor Q1. More specifically, an intermittent operation is an example of the adverse effect.
In FIGS. 22A and 22B, Vce is the collector-emitter voltage of the switching transistor Q1, Ic is the collector current of Q1, Vbe is the base-emitter voltage of Q1, and Ib is the base current of Q1. A great amount of switching loss is produced, as represented by the hatched area in FIG. 22A, by the product of the collector-emitter voltage of the switching transistor Q1 and the excessive current which flows through the collector of the switching transistor Q1 when it turns on.
In general, the output of the high-voltage power supply is turned on/off not by turning on/off the input power but in response to a remote control signal given from the outside of the high-voltage power supply while maintaining the input power supply in the on-state. In this case, the high-voltage power supply is required to have a steep rising-up characteristic without overshoot. In the conventional power supply shown in FIG. 16, a transistor serving as a remote switch may be disposed between the base and the emitter of the switching transistor Q1 so that the transistor is remote-controlled in response to an external signal. In this case, at the instant at which the remote-switching transistor turns off in response to a starting signal given from the outside, a starting current is supplied from the DC power circuit 11 to the base of the switching transistor Q1 via the starting resistor R1 and an oscillation starts. At the instant at which the oscillation starts, the output voltage on the secondary side is equal to 0 V and the phototransistor of the photocoupler PC is in an open state. Therefore, at the instant immediately after the start of oscillation, the charging time constant is determined by the resistor R5 and the capacitor C3. As a result, the on-period of the switching transistor Q1 has a maximum value at the instant at which the oscillation starts. Because this maximum value of the on-period is much greater than the rated value in the normal state, an initial voltage corresponding to the maximum on-period is induced across the secondary winding Ls and thus a great overshoot occurs.
A technique widely used to prevent the above problems in the conventional high-voltage power supply is to control the DC input voltage applied to the primary winding of the high-voltage transformer, as shown in FIG. 23, so as to obtain a stable output, instead of controlling the on-period of the switching transistor Q1. In FIG. 23, Q5 is a control power transistor which reduces the voltage of a DC input power supply 1 thereby controlling the input voltage applied to a high-voltage transformer T. In response to the signal detected by an output voltage detection circuit, a controller controls the base current of the transistor Q5 so as to obtain a stabilized output voltage. The switching transistor Q1 is periodically turned on and off at fixed time intervals by an oscillator.
However, because the switching power supply shown in FIG. 23 is based on the externally excited switching circuit, not only an external oscillator is required but also an additional power transistor for generating a reduced voltage input to the high-voltage transformer is required. Thus, the circuit becomes complicated in configuration and large in size.